Conventional high-frequency antennas are often cumbersome to manufacture. In particular, conventional beamforming antenna arrays require complicated feed structures and phase-shifters that are impractical to be implemented in a semiconductor-based design due to its cost, power consumption and deficiency in electrical characteristics such as insertion loss and quantization noise levels. In addition, such beamforming arrays become incompatible with digital signal processing techniques as the operating frequency is increased. For example, at the higher data rates enabled by high frequency operation, multipath fading and cross-interference becomes a serious issue. Adaptive beamforming techniques are known to combat these problems. But adaptive beamforming for transmission at 10 GHz or higher frequencies requires massively parallel utilization of A/D and D/A converters.
To address the need in the art for improved beamforming antenna arrays, the present inventor disclosed beamforming antenna arrays compatible with wafer scale integration in U.S. Ser. No. 11/074,027, filed Mar. 7, 2005, now U.S. Pat. No. 7,126,542. and U.S. Ser. No. 11/049,098, filed Feb. 2, 2005, now U.S. Pat. No. 7,126,541. the contents of both of which are hereby incorporated by reference. These applications utilized and expanded upon the beam forming capabilities disclosed by the present inventor in U.S. Ser. No. 10/423,303, filed Apr. 25, 2003, now U.S. Pat. No. 6,885,344, U.S. Ser. No. 10/423,160, filed Apr. 25, 2003, now U.S. Pat. No. 6,870,503 and U.S. Ser. No. 10/422,907, filed Apr. 25, 2003, Ser. No. 10/423,129, filed Apr. 25, 2003, now U.S. Pat. No. 6,963,307, Ser. No. 10/860,526, filed Jun. 3, 2004, now U.S. Pat. No. 6,982,670, and Ser. No. 10/942,383, filed Sep. 16, 2004, the contents of all of which are hereby incorporated by reference in their entirety.
One embodiment of a beamforming antenna system described in the above-mentioned applications is shown in FIG. 1, which illustrates an integrated RF beamforming and controller unit 130. In this embodiment, the receive and transmit antenna arrays are the same such that each antenna 170 functions to both transmit and receive. A plurality of integrated antenna circuits 125 each includes an RF beamforming interface circuit 160 and receive/transmit antenna 170. RF beamforming interface circuit 160 adjusts the phase and/or the amplitude of the received and transmitted RF signal responsive to control from a controller/phase manager circuit 190. Although illustrated having a one-to-one relationship between beamforming interface circuits 160 and antennas 170, it will be appreciated, however, that an integrated antenna circuit 125 may include a plurality of antennas all driven by RF beamforming interface circuit 160.
A circuit diagram for an exemplary embodiment of RF beamforming interface circuit 160 is shown in FIG. 2. Note that the beamforming performed by beamforming circuits 160 may be performed using either phase shifting, amplitude variation, or a combination of both phase shifting and amplitude variation. Accordingly, RF beamforming interface circuit 160 is shown including both a variable phase shifter 200 and a variable attenuator 205. It will be appreciated, however, that the inclusion of either phase shifter 200 or attenuator 205 will depend upon the type of beamforming being performed. To provide a compact design, RF beamforming circuit may include RF switches/multiplexers 210, 215, 220, and 225 so that phase shifter 200 and attenuator 205 may be used in either a receive or transmit configuration. For example, in a receive configuration RF switch 215 routes the received RF signal to a low noise amplifier 221. The resulting amplified signal is then routed by switch 220 to phase shifter 200 and/or attenuator 205. The phase shifting and/or attenuation provided by phase shifter 200 and attenuator 205 are under the control of controller/phase manager circuit 190. The resulting shifted signal routes through RF switch 225 to RF switch 210. RF switch 210 then routes the signal to IF processing circuitry (not illustrated).
In a transmit configuration, the RF signal received from IF processing circuitry (alternatively, a direct down-conversion architecture may be used to provide the RF signal) routes through RF switch 210 to RF switch 220, which in turn routes the RF signal to phase shifter 200 and/or attenuator 205. The resulting shifted signal is then routed through RF switch 225 to a power amplifier 230. The amplified RF signal then routes through RF switch 215 to antenna 170 (FIG. 1). It will be appreciated, however, that different configurations of switches may be implemented to provide this use of a single set of phase-shifter 200 and/or attenuator 205 in both the receive and transmit configuration. In addition, alternate embodiments of RF beamforming interface circuit 160 may be constructed not including switches 210, 220, and 225 such that the receive and transmit paths do not share phase shifter 200 and/or attenuator 205. In such embodiments, RF beamforming interface circuit 160 would include separate phase-shifters and/or attenuators for the receive and transmit paths.
To assist the beamforming capability, a power detector 250 functions as a received signal strength indicator to measure the power in the received RF signal. For example, power detector 250 may comprise a calibrated envelope detector. Referring back to FIG. 1, a power manager 150 may detect the peak power determined by the various power detectors 250 within each integrated antenna circuit 125. The integrated antenna circuit 125 having the peak detected power may be denoted as the “master” integrated antenna circuit. Power manager 150 may then determine the relative delays for the envelopes for the RF signals from the remaining integrated antenna circuits 125 with respect to the envelope for the master integrated antenna circuit 125. To transmit in the same direction as this received RF signal, controller/phase manager 190 may determine the phases corresponding to these detected delays and command the transmitted phase shifts/attenuations accordingly. Alternatively, a desired receive or transmit beamforming direction may simply be commanded by controller/phase manager 190 rather than derived from a received signal. In such embodiment, power manager 150 and power detectors 250 need not be included since phasing information will not be derived from a received RF signal.
Regardless of whether integrated antenna circuits 125 perform their beamforming using phase shifting and/or amplitude variation, the shifting and/or variation is performed on the RF signal received either from the IF stage (in a transmit mode) or from its antenna 170 (in a receive mode). By performing the beamforming directly in the RF domain as discussed with respect to FIGS. 1 and 2, substantial savings are introduced over a system that performs its beamforming in the IF or baseband domain. Such IF or baseband systems must include A/D converters for each RF channel being processed. In contrast, the system shown in FIG. 1 may supply a combined RF signal from an adder 140. From an IF standpoint, it is just processing a single RF channel for the system of FIG. 1, thereby requiring just a single A/D. Accordingly, the following discussion will assume that the beamforming is performed in the RF domain. The injection of phase and/or attenuation control signals by controller/phase manager circuit 190 into each integrated antenna circuit 125 may be performed inductively as discussed in U.S. Pat. No. 6,963,307.
Examination of FIG. 1 shows that a network is necessary for the distribution of the RF signals to and from the IF stage to integrated antenna units 125 as well as to and from RF beamforming interface circuits 160 and their corresponding antenna(s) 170. U.S. Pat. No. 7,126,542 discloses a micro-waveguide network for distributing these RF signals. Because of the use of waveguide transmission, very low transmissions losses were thereby introduced into the distributed RF signals. Moreover, the micro-waveguide network was compatible with wafer scale integration of the resulting beamforming array.
A wafer scale integrated antenna module (WSAM) as disclosed in U.S. Pat. No. 7,126,542 may include three primary layers. The first layer would be a semiconductor substrate, such as Si. On a first surface of the substrate, antennas such as patches for the integrated antenna circuits are formed as discussed, for example, in U.S. Pat. No. 6,870,503. Active circuitry for the corresponding integrated antenna circuits that incorporate these antennas on formed on a second opposing surface of the substrate. A micro-waveguide transmission network such as a rectangular waveguide network is formed adjacent this second opposing surface. The second layer would include the antennas on the first side of the substrate whereas the third layer would include the rectangular waveguide network. Thus, the WSAM includes the “back side” feature disclosed in U.S. Ser. No. 10/942,383 in that the active circuitry and the antennas are separated on either side of the substrate. In this fashion, electrical isolation between the active circuitry and the antenna elements is enhanced. Moreover, the ability to couple signals to and from the active circuitry is also enhanced.
Adjacent to the opposing second surface is the micro-waveguide distribution network. This network carries the RF signals to and from the antennas as discussed above. Thus, the network also distributes RF signals to and from the IF processing stage (or direct down-conversion stage depending upon the receiver architecture).
The network comprises waveguides that may be formed using metal layers in a semiconductor process such as CMOS as discussed in, for example, U.S. Ser. No. 10/423,160. However, it will be appreciated the waveguide diameter is then limited by maximum separation achievable between metal layers in such semiconductor processes. Typically, the maximum achievable waveguide diameter would thus be 7 microns or less, thereby limiting use of the waveguide to frequencies above 40 GHz. To accommodate lower frequency operation, micro-machined waveguides may also be utilized.
As discussed in U.S. Ser. No. 10/942,383, a heavily doped deep conductive junction through the substrate couples the active circuitry to vias/rods at the first substrate surface that in turn couple to the antenna elements. Formation of the junctions is similar to a deep diffusion junction process used for the manufacturing of double diffused CMOS (DMOS) or high voltage devices. It provides a region of low resistive signal path to minimize insertion loss to the antenna elements.
Upon formation of the junctions in the substrate, the active circuitry may be formed using standard semiconductor processes. The active circuitry may then be passivated by applying a low temperature deposited porous SiOx and a thin layer of nitridized oxide (SixOyNz) as a final layer of passivation. Thickness of these sealing layers may range from a fraction of a micron to a few microns. The opposing second surface may then be coated with a thermally conductive material and taped to a plastic adhesive holder to flip the substrate to expose the first surface. The substrate may then be back ground to reduce its thickness to a few hundreds of micro-meters.
An electric shield may then be sputtered or alternatively coated using conductive paints on background surface. A shield layer over the electric field may form a reflective plane for directivity and also shields the antenna elements. In addition, parts of the shield form ohmic contacts to the junctions. For example, metallic lumps may be deposited on the junctions. These lumps ease penetration of the via/rods to form ohmic contacts with the active circuitry.
The network may be formed in a glass, metallic, oxide, or plastic-based insulating layer. Depending upon the desired propagation frequency in the network, the thickness of the substrate may range from a few millimeters to multiple tens of microns. A rectangular or circular cavity is then etched into the insulating layer to form a waveguide cavity. The walls of the cavity may then be metallically coated using silver, copper, aluminum, or gold to provide the waveguide boundaries. Each integrated antenna circuit (FIGS. 1 and 2) will need a feedline/receptor to couple to the network as discussed, for example, in U.S. Pat. No. 7,126,541. Each feedline/receptor may be formed from a discrete metallic part such as a base pin that is inserted into the metallic lumps described above to form ohmic contacts to the active circuitry analogous to the insertion of the rods/vias. A metallic plate may then be used to seal the waveguide cavities to complete the micro-waveguide network. Because the network is metallic, it also may function as a heat sink for cooling the active circuitry.
Although this WSAM advantageously suffers relatively very little loss in signal propagation through the micro-waveguide network, the antenna array capacity is impacted by the relative size necessary for each waveguide chamber. In general, such chambers need to be approximately ½ wavelength across as known in the waveguide arts. In turn, however, this minimum width requirement limits the number of antennas that may be integrated into a single wafer as each antenna would require (ultimately) its own waveguide.
To address the need in the art for wafer scale beamforming antenna arrays that have improved array density, U.S. Ser. No. 11/141,283 discloses a beamforming antenna array in which the micro-waveguide network disclosed in U.S. Pat. No. 7,126,542 is replaced by either a coplanar or microstrip waveguide network. In this fashion, the pitch between adjacent conductors in the network is substantially reduced. However, the accompanying transmission losses are thereby substantially increased. For example, consider the losses at 40 GHz. A rectangular waveguide (such as disclosed in U.S. Pat. No. 7,126,542) has losses of less than 0.001 dB/mm at this frequency. However, microstrip and coplanar waveguides (CPWs) have losses of approximately 10 to considerably more than 100 times greater. For example, a thin field CPW network has losses of above 0.1 dB/mm at this frequency. To obtain the array density benefits of a CPW or microstrip network while obtaining satisfactory power transfer through the network, U.S. Ser. No. 11/141,283 discloses distributed amplifiers that may be integrated with the network.
Although the beamforming antenna array disclosed in U.S. Ser. No. 11/141,283 thereby obtains a high-density antenna array with sufficient gain provided by the distributed amplifiers, phase shifters are required to provide beam steering capabilities. Beam steering using electronic phase shifters has several advantages over mechanically-steered antennas. Phased array antennas are lighter in weight, more agile, and induce no angular momentum to a satellite as would be the case for a mechanically-steered antenna as the beam is redirected. Phased array antennas have traditionally used MMIC (Monolithic Microwave Integrated Circuits) phase shifters, which have two shortcomings: they can contribute substantially to the cost of fabrication and also tend to introduce a relatively high insertion loss. MEMS-controlled phase shifters have recently been demonstrated that show promise in lowering MMIC-induced cost and insertion loss. In addition, MEMS-based phase shifters have lower parasitics and higher linearity. However, a MEMS-based approach suffers from a lack of integration with the state-of-art Si technology.
Accordingly, there is a need in the art for improved phased array antenna architectures.